Electronic control systems and methods

ABSTRACT

An apparatus in an electronic control system allows two or three wire operation. A power supply can supply power to the enclosed circuitry in both two and three wire installations. Two separate zero cross detectors are used such that timing information can be collected in both two and three wire installations. Both zero cross detectors are monitored and are used to automatically configure the electronic control. Over voltage circuitry senses an over voltage condition across a MOSFET which is in the off state and turns the MOSFET on so that it desirably will not reach the avalanche region. Over current circuitry senses when the current through the MOSFETs has exceeded a predetermined current threshold and then turns the MOSFETs off so they do not exceed the MOSFETs&#39; safe operating area (SOA) curve. Latching circuitry is employed to keep the protection circuitry in effect even after a fault condition has cleared. Lockout circuitry is used to prevent one protection circuit from tripping after the other circuit has already tripped from a fault condition. The protection circuitry output is desirably configured such that it can bypass and override the normal turn on and turn off impedance and act virtually directly on the gates of the MOSFETs. Preferably, the system has a high efficiency switching type power supply in parallel with a low frequency controllably conductive device.

CROSS-REFERENCE TO RELATED APPLICATIONS

[0001] This application claims the benefit of U.S. ProvisionalApplication No. 60/303,508, filed Jul. 6, 2001, the entire disclosure ofwhich is hereby incorporated by reference.

FIELD OF THE INVENTION

[0002] The present invention relates in general to electronic controlcircuits and systems, and more particularly, to lighting controlcircuits and systems.

BACKGROUND OF THE INVENTION

[0003] There are many applications where it is desirable to control theamount of average electrical power delivered to a load. One example ofsuch an application is the use of a lighting dimmer to control theoutput of a lamp. A dimmer typically functions by controlling theconduction of current through the load. A controllably conductive deviceis synchronized to the AC line voltage and is controlled to conduct fora predetermined interval in each half cycle of the AC line voltage. Thatis, the load only receives power (is on) for a portion of the AC linevoltage half cycle. The longer the conduction time, the more powerdelivered to the load. By the same logic, the shorter the conductiontime, the less power delivered to the load.

[0004] There are primarily two methods for controlling AC loads such aslighting loads, forward phase control and reverse phase control. Acontrollably conductive device is a device whose conduction can becontrolled by an external signal. These include devices such as metaloxide semi-conductor field effect transistors (MOSFET), insulated gatebi-polar transistors (IGBT), bi-polar junction transistors (BJT),triacs, silicon controlled rectifiers (SCRs), relays, switches, vacuumtubes and the like. These two control methods utilize the conductive andnon-conductive states of a controllably conductive device to control thepower in a load and synchronize the conduction and non-conduction of thecontrollably conductive devices to zero crosses of the source of AC linevoltage.

[0005] The method of forward phase control, as shown in FIG. 13,synchronizes a controllably conductive device to the source of AC powerand controls the controllably conductive device to be non-conductiveover the first portion of an AC line voltage half cycle, then controlsthe controllably conductive device to be conductive over the remainingportion of the AC line voltage half cycle. In the method of reversephase control, as shown in FIG. 14, the periods of non-conduction andconduction are reversed with respect to time. That is to say, thecontrollably conductive device is controlled to be conductive during thefirst portion of the AC line voltage half cycle followed by a period ofnon-conduction in the same half cycle. The method of reverse phasecontrol is often used for operation of capacitive loads such aselectronic transformers.

[0006] In forward phase control based control systems the controllablyconductive device is often a triac or an SCR. These devices can becontrolled to be non-conductive or conductive. However, if they arecontrolled to be conductive, they can only be made non-conductive byallowing the current through them to go to zero. Due to thischaracteristic, these types of controllably conductive devices are notused for reverse phase control based control systems where the abilityto enable and disable conduction is required.

[0007] Electronic controls need to derive a power supply in order topower their associated electronics. Additionally, many controls requireline frequency related timing information. Controls which only have twopower terminals have one of these terminals (the hot terminal) connectedto a hot wire of a source of AC power and the other terminal (the dimmedhot terminal) connected to a first terminal of a load. Controls withthis type of connection are often referred to as “two wire” controls.Two wire controls which are connected in series with their loads mustcharge their power supplies and obtain timing information through thisload. The load can often have a wide range of input impedance. As such,the operation of the power supply and timing circuit is oftencompromised in the two wire connection scheme. However, a two wireconnection is necessary when the control is wired in an applicationwhere a neutral wire is not available.

[0008] Controls which have connections to the hot wire, load, andneutral wire are often referred to as “three wire” controls. When aneutral wire from the source of AC power is available for connection toa neutral terminal of the control, the power supply and zero crossinformation can be derived independently of the connected load, therebyenhancing performance. In many applications, a neutral wire from thesource of AC power is not available. Therefore, a control is needed thatcan operate correctly as either a two wire or three wire control,thereby allowing the control to be used in a broad range of fieldapplications with great flexibility.

[0009] Prior art for developing a non-isolated low voltage power supplyfrom a high voltage source, such as the AC line voltage, used circuitssuch as a cat ear power supply. Such a system would conduct at or nearthe line voltage zero cross so as to recharge an energy storagecapacitor. Such systems typically operate properly in the region about 1millisecond from the zero crossing of the line voltage. Operationoutside that time window can cause excessive power to be dissipated inthe power supply.

[0010] The cat ear power supply has relatively high peak and highaverage input currents with respect to the average current supplied tothe connected DC load. This high average input current presents asignificant problem when this supply technology is used with electroniclow voltage (ELV) load types on phase control dimmers connected in a twowire mode. A supply for low voltage control circuitry is needed that haslow average input currents through the high voltage load. Also, typicalprior art power supplies have been relatively inefficient so that theyrequire higher average input currents to supply the power requirementsof typical prior art dimmers.

[0011] Another disadvantage of prior art power supplies for lightingcontrol devices is that power losses in the power supplies increase withthe amount of current required to be delivered by the power supply. Thetrend in modern lighting controls is to incorporate more features andfunctionality. These features and functionality require ever increasingamounts of current to be delivered by the power supply. Hence, it isdesired to provide a power supply for a lighting control able toefficiently supply greater amounts of current than are presentlyavailable from typical prior art power supplies without the power lossesassociated with such prior art power supplies.

[0012] There are a variety of fault conditions to which lightingcontrols may be subject, including, for example, over voltage and overcurrent conditions. Over voltage conditions can be caused by, forexample, the turning on and off of nearby and connected magnetic loads,capacitive coupling to parallel wire runs with sharp transient loads,lightning strikes, etc. Over current conditions can be caused by, forexample, short circuited loads, connected loads exceeding the controlsrating, mis-wire conditions, etc. Semiconductor devices, such asMOSFETs, have limits as to how much voltage and current they canwithstand without failure. In order to protect a control that uses thesesemiconductor devices from failure, these limits are preferably neverexceeded. Fast detection of fault conditions, and fast reaction theretois desirable in order to protect these devices.

[0013] In contrast, during normal operation, the rates of transitionbetween conductive and non-conductive states of these semiconductordevices are controlled to be slow. These slow rates of transition areused, for example, to limit the voltage and current waveforms as seen bythe load, to comply with radiated and conducted radio frequencyinterference (RFI) limits, or to limit voltage ringing caused byinductive power wiring. However, these slow rates of transition duringnormal operation are too slow for adequate protection of thesesemiconductor devices. Thus, there is a need for protection circuitrythat operates to cause fast rates of transition under fault conditions,while still allowing these semiconductor devices to be operated withslow rates of transition under normal operating conditions.

SUMMARY OF THE PRESENT INVENTION

[0014] The present invention is directed towards an apparatus in anelectronic control system which will allow two or three wire operation.According to aspects of the invention, the apparatus employs a highefficiency power supply which can supply power to the operatingcircuitry of the electronic control system in both two and three wireinstallations.

[0015] According to another aspect of the invention, the apparatusemploys a detector that detects the presence of a neutral wireconnection and outputs a signal responsive to the detected neutral wireconnection to cause the electronic control system to operate in a twowire mode when the neutral wire connection is absent and to operate in athree wire mode when the neutral wire connection is present.

[0016] According to other aspects of the invention, the apparatusemploys a zero cross detector which can operate in both two and threewire modes. In an embodiment, the zero cross detector comprises a hotzero cross detector that generates a hot zero cross signal and a neutralzero cross detector that generates a neutral zero cross signal, and amicroprocessor responsive to the zero cross signals to cause theapparatus to operate in one of the two wire and three wire modes.

[0017] According to yet another aspect of the invention, the apparatusemploys a system for stabilizing the zero crossing signal received bythe electronic control system when the system is operating electroniclow voltage transformer connected loads.

[0018] Another embodiment of the present invention is directed towardthe protection of controllably conductive devices such as semiconductordevices like MOSFETs and IGBTs used in an electronic control system.Over voltage circuitry senses an over voltage condition across acontrollably conductive device which is in the non-conductive state andcontrols the controllably conductive device to be conductive so as toremove the over voltage condition. Over current circuitry senses whenthe current through a controllably conductive device has exceeded apredetermined current threshold and controls the controllably conductivedevice to be non-conductive so as to ensure the safe operating area ofthe controllably conductive device is not exceeded. The protectioncircuitry output is desirably configured such that it can bypass andoverride the normal control path of the controllably conductive deviceand cause the controllably conductive device to transition rapidlybetween conduction and non-conduction states.

[0019] According to further aspects of the invention, latching circuitryis employed to keep the results of the protection circuitry in effecteven after the fault condition has cleared. Lockout circuitry is used toprevent one protection circuit from tripping after the other protectioncircuit has already tripped from a particular fault condition.

[0020] The foregoing and other aspects of the present invention willbecome apparent from the following detailed description of the inventionwhen considered in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

[0021] For the purpose of illustrating the invention, there is shown inthe drawings an embodiment that is presently preferred, it beingunderstood, however, that the invention is not limited to the specificmethods and instrumentalities disclosed. In the drawings:

[0022]FIG. 1 is a high level block diagram of an exemplary system inaccordance with the present invention;

[0023]FIG. 2 is a block diagram of an exemplary control system inaccordance with the present invention;

[0024]FIG. 3 is a circuit schematic diagram of a portion of an exemplarycontrol system in accordance with the present invention;

[0025]FIG. 4 is a circuit schematic diagram of another portion of anexemplary control system in accordance with the present invention;

[0026]FIG. 5 is a circuit schematic diagram of another portion of anexemplary control system in accordance with the present invention;

[0027]FIG. 6 is a circuit schematic diagram of another portion of anexemplary control system in accordance with the present invention;

[0028]FIG. 7 is a simplified block diagram of an exemplary transistordriver in accordance with the present invention;

[0029]FIG. 8 is a simplified block diagram of an exemplary zero crossdetector in accordance with the present invention;

[0030]FIG. 9 is a simplified schematic diagram of an exemplary steeringcircuit in accordance with the present invention;

[0031]FIG. 10 is a simplified schematic diagram of an exemplary systemused for eliminating false indication of zero crossing in accordancewith the present invention along with exemplary timing diagrams;

[0032]FIG. 11 is a circuit schematic diagram of an exemplary load foruse with the present invention;

[0033]FIG. 12 is a block diagram of an exemplary system comprising a lowvoltage power supply in parallel to a high voltage controllablyconductive device in accordance with the present invention;

[0034]FIG. 13 is a diagram which illustrates an exemplary forward phasecontrol waveform; and

[0035]FIG. 14 is a diagram which illustrates an exemplary reverse phasecontrol waveform.

DESCRIPTION OF EXEMPLARY EMBODIMENTS AND BEST MODE

[0036] An embodiment of the present invention is directed to anelectronic control system and in particular a lighting controller thatcan automatically determine whether to operate in two wire mode or threewire mode (i.e., to operate with or without a neutral wire connection).The controller senses whether there is a neutral wire connection to theelectronic control system and adjusts its operation accordingly. Theelectronic control system automatically selects and continuouslymonitors the connection scheme. An embodiment is directed toward anelectronic control system such as a lighting controller or dimmer;however, the invention has broader application in other electroniccontrols.

[0037]FIG. 1 is a high level block diagram of an exemplary system inaccordance with the present invention. An electronic control system 100,also referred to herein as a lighting controller or a dimmer, ispreferably connected between an input source, such as an AC linevoltage, and a first terminal of a load 200, such as an incandescentlamp or an electronic low voltage (ELV) transformer with a connectedlamp load. A typical AC line voltage comprises a 120 volt, 60 Hz, singlephase power source. The AC line may also comprise a 220 to 240 volt, 50Hz single phase power source, or the like.

[0038] The electronic control system 100 comprises a hot terminal, adimmed hot terminal, and a neutral terminal which is optionallyconnected to the neutral wire of the AC line. The neutral wire of the ACline is also connected to a second terminal of the load 200.

[0039] The electronic control system 100 controls the flow of current tothe load 200 using either forward phase control or reverse phase controlbased on a predetermined selection. For electronic low voltage loads, itis desirable to operate with reverse phase control because electroniclow voltage loads have a capacitive input impedance. If forward phasecontrol is used to control electronic low voltage loads, a largetransient current can flow when the controllably conductive device ofthe electronic control system transitions from a non-conductive to aconductive state.

[0040] The electronic control system 100 detects whether the neutralwire is connected and adjusts its operation accordingly. In particular,as described in further detail below a microprocessor monitors theoutput of a detector, and determines which of two wire or three wiremode should be used by the electronic control system to control theconnected load.

[0041]FIG. 2 is a block diagram of an exemplary electronic controlsystem 100, and FIGS. 3, 4, 5, and 6 are circuit schematic diagrams ofvarious portions of an exemplary electronic control system 100. Theelectronic control system 100 comprises a zero cross detector 110, anover voltage protection circuit 120, an over current protection circuit130, a power supply 150, output circuit 160, and a microprocessor 190.The hot terminal and neutral terminal are connected to the zero crossdetector 110, and the dimmed hot terminal is provided to the overvoltage protection circuit 120.

[0042] The power supply 150 is preferably a switching power supply withhigh efficiency (e.g., an efficiency above about 50%). Moreparticularly, with respect to FIG. 3, the power supply 150 is suppliedwith sufficient energy in both the two wire mode and three wire mode.The diodes D1, D2, D60, D61, and the two body diodes of the MOSFETs Q101and Q102 (in output circuit 160 shown in FIG. 5) form a full wave bridgefor power supply current to flow in both AC line voltage half cycles.

[0043] In the case of an electronic control system with the neutralterminal connected to the neutral wire of the AC line voltage (threewire mode), the bus capacitor C10 of the power supply 150 charges bydrawing current from the source of AC power through the hot wire andneutral wire in the negative half cycle of the AC line voltage andthrough the hot wire and the load in the positive half cycle of the ACline voltage. In the case of a two wire mode, the bus capacitor C10 ischarged in both half cycles through the load when the absolute value ofthe AC line voltage is greater than the bus capacitor voltage V_(BUS)and the controllably conductive devices are non-conductive. The diodeD10 of FIG. 3 prevents the bus capacitor C10 from discharging throughother connected circuitry. The bus capacitor C10 is used as a source ofhigh voltage DC to power an efficient power converter to provide lowvoltage DC to operate the control circuits of the electronic controlsystem.

[0044] The efficient power converter operates as follows using the wellknown buck converter topology. The efficient power converter includesthe following principal components U10, L10, C13, and a regulationcircuit including principal components U11, Z10, and R12. When thevoltage across capacitor C13 is below the voltage threshold determinedby the series combination of zener Z10 and the LED diode drop ofoptocoupler U11, current will not flow through those components, thusthe opto-coupled transistor of optocoupler U11 will be off. When thetransistor is off, no current can flow from the enable pin 4 ofcontroller U10 (such as, for example, a TNY253 IC manufactured by PowerIntegrations, Inc. San Jose, Calif.) to its source pin 2,3, therebyenabling controller U10 to begin switching in order to raise the outputvoltage level of C13. The controller U10 will then turn on its internalMOSFET, thereby allowing current to flow from the drain to the source,through the inductor L10 and into the output capacitor C13. The rate ofrise of this current is limited by the inductance of inductor L10. Whenthe current in the internal MOSFET reaches the internally set thresholdof controller U10, the internal MOSFET is turned off. The current willcontinue to flow around the loop defined by inductor L10, capacitor C13,and diode D11 until the current in the inductor reaches zero. Thisswitching cycle is repeated at a maximum rate of 44 kHz as set bycontroller U10, until the voltage across capacitor C13 exceeds thevoltage threshold determined by the series combination of zener Z10 andthe LED diode drop of optocoupler U11. When this voltage threshold isexceeded, current will begin to flow through those components, therebyturning the opto-coupled transistor of optocoupler U11 on. When thistransistor turns on, the enable pin 4 of controller U10 is therebyconnected to the source pin 3, and in accordance with the operation ofcontroller U10, switching is terminated. Additionally, the enable pin 4can be used to select a running or non-running mode of the power supply.This pin can be used to constrain the operation of the power supply toselected times of the AC line voltage half cycle. Since switch modepower supplies generate electrical noise, it is advantageous toconstrain the operation of the power supply to times when other noisesensitive circuits are not operating.

[0045] In prior electronic control systems which include a power supplyutilizing a high frequency switching converter, the power supply isconnected to draw current directly from a low impedance source such asan AC line voltage. In the apparatus of an embodiment of the invention,the power supply, utilizing a high frequency switching converter, drawscurrent through the load which may typically have a high impedance.

[0046] It is desirable to provide an over voltage protection circuit 120and an over current protection circuit 130 that will sense and react toan over voltage across or an over current condition through acontrollably conductive device in an electronic control system toprotect the electronic control system from damage.

[0047] Circuit details of an exemplary over voltage protection circuit120 and an exemplary over current protection circuit 130 are shown inFIG. 4. At startup, a reference voltage V_(REF) for the comparatorsU110:A, U110:B is derived from the 8 V MOSFET drive rail, Vc, throughthe current limiting resistor R114, voltage regulating zener Z111, and anoise decoupling capacitor C111. It is desirable to power thecomparators in IC U110 with 8V as opposed to 5V to allow the use of asharp-knee 5.6V zener as the reference voltage to which the detectioncircuits are compared. A well regulated voltage reference tightens thetolerancing window on the detection circuits.

[0048]FIG. 7 contains a simplified block diagram of an exemplary outputcircuit. Circuit details of an exemplary output circuit 160 are shown inFIG. 5. It is well known that the rate of transition between the statesof conduction for a MOSFET can be controlled by selecting the impedanceof the drive circuit. The higher the impedance the slower the transitionrate. The output transistors Q101 and Q102 are driven through highimpedance path 165, during normal operation, and through low impedancepath 162 (FIG. 4) during a fault condition. The microprocessor 190 isconnected to the high impedance path 165 and the protection circuits120, 130. The protection circuits 120, 130 are also connected to the lowimpedance path 162. When the protection circuits 120, 130 detect afault, the low impedance path 162 is activated. The low impedance path162 is only active when a fault is detected. The fault path overridesthe normal path provided by the high impedance path 165.

[0049] In normal operation, the high impedance path 165 is used. Thetransistors Q101 and Q102 are turned on through resistors R103 and R104,and are turned off through resistor R104. During normal operation,transistor control is provided by two microprocessor ports, Gate Driveand Gate Drive Complement (shown in FIG. 6). To turn on the MOSFETs Q101and Q102, Gate Drive is driven high, thereby turning on transistorQ100:B (shown in FIG. 5), thereby turning on transistor Q100:A, whichapplies 8V to the gates of MOSFETs Q101 and Q102 through a resistanceset by the series combination of resistors R103 and R104. When GateDrive is high, Gate Drive Complement is low thereby turning offtransistor Q123:B, thus opening the current path from 8V to circuitcommon.

[0050] To turn off the MOSFETs Q101 and Q102, Gate Drive is pulled low,thereby turning transistor off Q100:B, thereby turning off transistorQ100:A, opening the current path from the 8V rail to the gates ofMOSFETs Q101 and Q102 gates. Gate Drive Complement is driven high,turning on transistor Q123:B, thereby discharging the gates of MOSFETsQ101 and Q102 through the resistor R104.

[0051] The MOSFETs Q101 and Q102 get driven through the high impedancepath to reduce RFI emissions during normal operation. During a faultcondition, the MOSFETs Q101 and Q102 are driven through the lowimpedance path to shut them down quickly.

[0052] During normal operation, the voltage on the inverting inputterminal of comparator U110:A (the Over voltage protection circuit (OVP)comparator) is less than the reference voltage of 5.6V so the output ofthis comparator U110:A is high impedance. This high impedance will keepthe transistor Q111:A off and the MOSFETs Q101 and Q102 are unaffected.The microprocessor port OVP_RESET (shown in FIG. 6) is low whenever theMOSFETs Q101 and Q102 are off, thereby turning off transistor Q111:B andenabling the detector.

[0053] Additionally, the reference voltage on the inverting terminal ofcomparator U110:B (the Over current protection circuit (OCP) comparator)is less than the 8V on the non-inverting terminal so the output of thiscomparator U110:B is high impedance and the MOSFETs Q101 and Q102 areunaffected. Diodes DN111:1 and DN120:1 provide isolation between theMOSFETs Q101 and Q102 and the protection circuitry 120, 130.

[0054] During an over voltage fault condition, as the voltage across theMOSFETs Q101 and Q102 rises so does the divided down voltage atresistors R110 and R111's common node. When this node's voltage, whichis also connected to comparator U110:A's inverting terminal, exceeds thereference voltage V_(REF), the output of the comparator U110:A will pulllow, thereby turning on transistor Q111:A, thereby applying drivevoltage to the gates of MOSFETs Q101 and Q102 via a low impedance pathset by resistor R129. The low impedance path allows the MOSFETs Q101 andQ102 to turn on at a faster rate than during the normal mode ofoperation. Because voltage transients can be on the order of severalthousand volts, the input voltage to the OVP comparator is safelyclamped by diode DN110:1 to a maximum of about 8.6V.

[0055] The OVP circuit 120 is latched on, even after the fault conditionis clear, by virtue of the feedback action of diode DN111:2. Thisfeedback keeps the inverting terminal 5 voltage of the comparator U110:Aabove the reference voltage V_(REF), thereby keeping transistor Q111:Aon.

[0056] The OVP latch is cleared by briefly driving the microprocessorport OVP_RESET high, thereby turning transistor Q111:B on and drivingpin 2 of comparator U110:A below the reference voltage V_(REF), therebydriving the output of the comparator U110:A to high impedance.

[0057] In order to prevent an oscillatory condition from occurringbetween over voltage protection and over current protection when oneprotection circuit trips, the other protection circuit is locked out.When over voltage protection circuit 120 activates, over currentprotection circuit 130 is disabled via diode DN120. The anode of DN120will be at approximately 7.4V when the over voltage protection circuit120 is activated, and this will hold the non-inverting terminal of theover current protection comparator U110:B high enough above thereference voltage V_(REF), even if the over current protection circuit130 tries to pull the non-inverting terminal low. This effectivelydisables the over current protection comparator U110:B.

[0058] During an over current fault condition, as the current throughthe MOSFETs increases, the voltage across resistor R109 (in outputcircuit 160) increases. As the voltage approaches 0.6V, eithertransistor Q120:A or Q120:B will begin to turn on depending on thedirection of current flow. The turn on of the transistors Q120:A, Q120:Bwill pull the non-inverting terminal of the comparator U110:B down belowthe reference voltage V_(REF), thereby switching the comparator's outputlow. This low output quickly turns off the MOSFETs Q101 and Q102 throughdiode DN120:1 and resistor R128. Noise filtering is provided byresistors R124 and R121, and capacitors C120, C121, and C122.

[0059] The over current protection circuit 130 is latched on, even afterthe fault condition has cleared, by virtue of the feedback action ofdiode DN120:2. This feedback keeps the non-inverting terminal of thecomparator U110:B below the reference voltage V_(REF), thereby keepingthe output low. The over current protection circuit is reset when GateDrive Complement goes high, turning on transistor Q123:B (in outputcircuit 160), which then turns on transistor Q123:A, thereby driving thenon-inverting terminal of the comparator U110:B to 8V and clearing thelatch.

[0060] When the over current protection circuit 130 activates, the overvoltage protection circuit 120 is disabled via diode DN110. When theoutput of the over current protection comparator U110:B goes low, theinverting terminal of the over voltage protection comparator U110:A ispulled to approximately 0.8V, thereby preventing the over voltageprotection circuit from activating.

[0061] The voltage comparators U110:A and U110:B provide fast reactionspeed and accuracy and work well across a wide temperature range. Eachcomparator has a specified typical response time of about 1.5 μsec withabout a 5 mV overdrive. The input offset voltage has a specified typicalvalue of about 2.0 mV at 25° C. The input to output response time of thecomparators with inputs driven to the rails is about 90 nanosec. In theover current protection circuit 130, the time from the input V_(REF)crossing to the MOSFET's 90% off point was measured to be about 3.5μsec. In the over voltage protection circuit 120, the time from theinput V_(REF) crossing to the MOSFET's 90% on point was measured to beabout 2.0 μsec.

[0062]FIG. 8 is a simplified block diagram of an exemplary zero crossdetector 110. The zero cross detector 110 comprises a hot zero crossdetector 112 that provides a hot zero cross detection signal and aneutral zero cross detector 115 that provides a neutral zero crossdetection signal when the neutral terminal is connected to a neutralwire. The microprocessor 190 monitors the output of the detectors 112and 115. If a neutral zero cross detection signal is sensed by themicroprocessor 190, it is determined that the connection is a three wireconnection and the three wire mode is activated in which the neutralzero cross detection signal from the neutral detector 115 is used fortiming. Otherwise, it is determined that the connection is a two wireconnection and the two wire mode is activated in which the hot zerocross detection signal from the hot detector 112 is used for timing.

[0063] Regarding the zero crossing detector 110, an example of which isshown in further detail in FIG. 3, generation of the hot zero crossdetection signal, which is used in the two wire mode, is accomplishedvia the hot zero cross detector 112 which is connected between the hotterminal and circuit common. Circuit common is connected to the dimmedhot terminal through the body diode of MOSFET Q102 and is connected tothe hot terminal through the body diode of MOSFET Q10l. Circuit commonwill have the same potential as the dimmed hot terminal during thepositive half cycle of the AC line voltage and will have the samepotential as the hot terminal during the negative half cycle of the ACline voltage. Resistors R63 and R64 divide down the voltage between hotand circuit common. When this divided down voltage reaches about 0.6V,transistor Q60:A will turn on, thereby pulling the normally logic highmicroprocessor port, HOT_ZC (shown in FIG. 6), to circuit common. Themicroprocessor senses this transition and thereby acquires the zerocross timing information. In detector 112, capacitor C61 is a noisedecoupling capacitor.

[0064] When the neutral terminal of the electronic control system isconnected to the neutral wire, it is desirable to acquire zero crosstiming information from the neutral zero cross detector which isconnected between the neutral terminal and hot terminal. Acquiring zerocross timing information in this manner is independent of the connectedload and is not subject to variations in the load which can cause zerocrossing time shifts particularly in the cases of magnetic or capacitiveloads. In addition, zero cross information can be acquired even when theelectronic control system is applying full line power to the load. Whenfull power is being delivered to the load 200, the hot zero crossdetector 112 does not produce a signal because the hot terminal and thedimmed hot terminal are at substantially the same potential and thusthere is substantially no voltage between the hot terminal and circuitcommon.

[0065] The neutral zero cross detector 115 creates transitions in thesame manner as the hot zero cross detector 112 but the output signalsare connected to the NEUT_ZC microprocessor port. The neutral zero crossdetector 115 employs two diodes that the hot zero cross detector 112does not: diode D60 protects the base emitter junction of transistorQ60:B from exceeding its rated reverse voltage by blocking current flowwhen circuit common is at the same potential as the hot terminal; anddiode D61 blocks current flow from the hot terminal, which wouldundesirably trigger the neutral zero cross detector 115 in the positivehalf cycle, when the MOSFETs Q101 and Q102 are non-conductive. Themicroprocessor 190 can be any type of microprocessor, such as a MotorolaMC68HC908AB32, as shown in FIG. 6.

[0066] The zero cross detector described above provides zero crosstiming information as well as neutral wire connection information to themicroprocessor. A separate neutral wire connection detector could beprovided which is separate from the zero cross detector described above.The primary function of a neutral wire connection detector is toindicate the presence of a neutral wire connection. The neutral wireconnection detector can provide information to the microprocessor as towhich of the two wire or three wire modes should be used. Other types ofneutral wire connection detectors may be used, such as mechanicaldetectors, in which a mechanical sensor detects the presence of aneutral wire and provides information to the microprocessor as to thestate of the neutral wire connection. A manual switch, or set ofswitches, such as a DIP switch could also be used to manually indicatethe presence of a neutral wire connection.

[0067]FIG. 9 is a simplified schematic diagram of an exemplary steeringcircuit that charges the bus capacitor C10 through the neutral terminalwhen a neutral wire is connected (e.g., in three wire mode). CapacitorC10 can be charged through multiple paths, from the hot terminal,neutral terminal, or dimmed hot terminal. The capacitor C10 is chargedfrom the hot terminal through diode D2, from the neutral terminalthrough diodes 60, 61, and through the dimmed hot terminal through diodeD1.

[0068] Typical prior art two wire electronic control systems control thepower delivered to a load by making the controllably conductive devicesconductive for a single selected portion of each AC line voltage halfcycle. Prior to the time of an expected zero crossing of the AC linevoltage, circuitry is enabled that opens a detection window to receive azero cross signal. When the zero crossing signal is received, theelectronic control system is synchronized to the AC line voltage andthus the conduction of the controllably conductive device issynchronized to the received zero crossing signal.

[0069] For an electronic control system operating in two wire mode, thiscontrol technique works well when the load impedance is primarilyresistive. When this technique is used with electronic low voltagelighting loads a problem arises due to the complex input impedance ofthe electronic low voltage transformer. Typical electronic low voltagetransformers operate by chopping the voltage applied to their inputterminals at a high frequency and stepping down the chopped voltagethrough a high frequency transformer. The circuitry to perform thischopping action operates in different modes depending on the inputvoltage to the electronic transformer. When the input voltage is low,typically less than about 60 volts, the chopper circuit is not runningand the input impedance of the transformer is very high and the inputcapacitor of the electronic transformer holds the actual value of thevoltage on the transformer when the chopping action ceases. When theline voltage reaches about 60 volts, the chopper circuit begins runningand the input impedance essentially drops to the impedance presented bythe connected lamp load. Additionally, during the period when thechopper is not running, the input capacitor is susceptible to chargingvia any leakage path through the electronic control system. Since theleakage currents are variable and based on multiple parameters, thecharging of the input capacitor of the electronic transformer is highlyvariable. This results in a variable voltage being present on the inputcapacitor of the electronic low voltage transformer at the start of anAC line voltage half cycle, effectively causing a variance in theinitial conditions for the operation of the electronic low voltagetransformer on a half cycle to half cycle basis. This variationinteracts with the zero crossing detection circuitry of a typical twowire phase control electronic control system, such as that of a lightingcontroller, so as to cause an instability in the zero crossing signal.This instability in the zero crossing signal introduces an instabilityin the conduction time of the controllably conductive devices and thus aflicker effect in the connected lamp load.

[0070] In order to stabilize the zero crossing signal available in twowire mode for an electronic control system operating an electronic lowvoltage transformer, it is necessary to stabilize the initial voltagecondition of the input capacitor of the electronic low voltagetransformer near the zero crossing of an AC line voltage half cycle. Ithas been found that this can be accomplished by allowing a very briefperiod of conduction to occur near the time of the zero crossing of theAC line voltage half cycle. In one embodiment, the controllablyconductive device in the electronic control system is controlled to beconductive for a duration of about 200 microseconds at a time about 1millisecond before an AC line voltage zero crossing. This brief periodof conduction when the AC line voltage is very low in absolute valueeffectively resets the input capacitor of the electronic low voltagetransformer to a consistent initial condition and therefore stabilizesthe zero crossing signal that is received in the electronic controlsystem.

[0071]FIG. 10 is a simplified block diagram of an exemplary circuit usedfor eliminating the instability of the zero crossing signal inaccordance with the present invention along with exemplary timingdiagrams.

[0072] For two wire operation, the transistors Q101 and Q102 of outputcircuit 160 are controlled to be conductive for a predetermined lengthof time at a predetermined point in time of each AC line voltage halfcycle, prior to a time when the microprocessor opens a zero crossingdetection window. For three wire operation, the transistors Q101 andQ102 preferably remain conductive through the time of AC line voltagezero crossing.

[0073] The load 200 (such as an electronic low voltage transformer shownin circuit schematic form in FIG. 11) is connected to the electroniccontrol system 100. The load 200 comprises capacitors C1, C2 that getcharged, and the voltage on these capacitors affects the operation ofthe electronic low voltage transformer operation and the zero crossingsignal received by the electronic control system 100. In two wire mode,the zero crossing of the AC line voltage is detected by measuring thevoltage drop across the dimmer (V_(DIMMER)) from the hot terminal to thedimmed hot terminal. However, when the MOSFETs Q101, Q102 are nonconductive, such as during the time preceding an AC line voltage zerocrossing, the voltage drop across the dimmer is equal to the AC linevoltage (V_(LINE)) minus the voltage drop across the load 200(V_(LOAD)). Because of leakage current through the dimmer, the capacitorC2 is able to charge toward some break-over voltage determined by thediac in the load 200. This causes the dimmer voltage V_(DIMMER) to belower than it would be otherwise. Undesirably, the load voltage V_(LOAD)may not be consistent from one zero crossing detection window to thenext, thereby causing the dimmer voltage V_(DIMMER) to be inconsistentfrom one zero crossing detection window to the next. This problem canmanifest itself to users as undesirable light flicker, especially at lowends when the lamp is dimmed.

[0074] Therefore, as previously discussed, to eliminate this problem intwo wire mode, the transistors Q101 and Q102 are controlled to beconductive (FET gate drive high) for a predetermined time period (e.g.,preferably at least about 200 μsec, and more preferably about 250 to 300μsec) and then controlled to be non-conductive before the start of thenext zero crossing detection window. The transistors Q101 and Q102 arecontrolled to be conductive at a line voltage sufficient to break-overthe diac in the load 200. The transistors Q101 and Q102 are controlledto be non-conductive prior to the start of the zero crossing detectionwindow. After the transistors Q101 and Q102 are controlled to benon-conductive, the microprocessor 190 opens or starts a zero crossingdetection window and begins monitoring the zero cross detector 110 forthe zero cross signal. Preferably, the zero crossing detection window isopened about 1 millisecond prior to where the zero cross signal isexpected and closed about 2 milliseconds after being opened.

[0075] The minimum duration for which the MOSFETs Q101, Q102 arecontrolled to be conductive for the purpose of eliminating theinstability of the zero crossing signal is determined by the desiredeffect upon a target set of electronic transformers for use with theelectronic control system 100. That is, the MOSFETs must be on for asufficient period of time, at a sufficiently high line voltage level, sothat the control circuits in the target set of electronic transformersbreak over into conduction, thereby causing the voltage across the loadto be returned to a consistent value from one zero crossing detectionwindow to the next. The maximum duration for which the MOSFETs Q101,Q102 are controlled to be conductive for the purpose of eliminating theinstability of the zero crossing signal is determined by many factors,such as the effect on visible light output from any lamp driven by theelectronic low voltage transformer, and switching and conduction lossesin the MOSFETs. For example, the longer the MOSFETs are allowed toremain in conduction, the more likely it is that current may flowthrough the load or that the light output may increase above a desiredlevel.

[0076] The microprocessor 190 monitors the line frequency and determineswhere the next zero crossing detection window will be opened.Preferably, the zero crossing detection window is opened at a time priorto the next expected AC line voltage zero crossing that is about 10% ofthe measured period of the AC line voltage half cycle. The advantageousstabilization of the zero cross signal described above can also improvethe operation of electronic control system operating in three wire modeby eliminating the effects of any leakage currents from the electroniccontrol system that flow through the electronic low voltage transformerthat may adversely effect the control circuits of the electronic controlsystem. Additionally, since in three wire mode operation of anelectronic control system the zero crossing signal is derived from thehot terminal and the neutral terminal, the controllably conductivedevices can remain conductive through the time of the AC line voltagezero crossing while achieving the above mentioned advantageous effectsof zero crossing stabilization.

[0077] Thus, for both two and three wire implementations, preferably,the zero crossing reference is reset regardless of the load. Thisprovides a clear consistent zero crossing reference.

[0078]FIG. 12 is a simplified schematic diagram of an exemplary highfrequency switching power supply in parallel with controllablyconductive devices Q110,Q102. The power supply 150 draws a low currentthrough the high voltage load 200 by using a switching converter toefficiently convert the high voltage across controllably conductivedevices Q101,Q102 to a low voltage supply voltage. The presentembodiment comprises a combination of a switching converter connected inparallel with a pair of high voltage controllably conductive devices.The MOSFETs Q101,Q102 in FIG. 12 represent the high voltage controllablyconductive devices. The gates of these devices Q101,Q102 are driven bycontrol circuitry powered by the low voltage power supply 150. Thiscombined system, in this case, is controlling one or more electronic lowvoltage transformers (load 200).

[0079] To further describe aspects of the invention, a conventionallinear regulator cat ear power supply used for prior art two wire modedimmers is typically about 10% efficient at converting power from a highvoltage source to a low voltage load (i.e., control circuitry), whereasthe power supply of the invention has an efficiency of about 75%. Forelectronic controls system requiring on the order of about 50 to 100 mWof power to operate their control circuits, about 0.5 to 1 watt of powerwould be dissipated in the power supply. In general, this has not been asignificant issue. However, associated with the low efficiency of thecat ear power supply are high peak and average input currents into thepower supply for a given average output current. Generally, the peakcurrent into a cat ear power supply is at least 10 times the averageoutput current. In the case of two wire mode dimmers, the peak currentdrawn by a cat ear power supply through the connected load can cause theload to make audible noise, particularly in the off state when the loadis expected to have no significant current flowing through it. The highaverage current of the cat ear power supply when directed through anelectronic low voltage transformer can cause flicker due to variationsin the zero cross signal as described above. Additionally, theefficiency of the cat ear power supply deteriorates as the differencebetween input voltage and output voltage increases. Therefore, operatingthe cat ear power supply beyond about the first 1 millisecond of the ACline voltage after a zero crossing is fundamental limitation. This limitof available conduction time for the cat ear power supplycorrespondingly causes the input peak current to rise significantly if asmall additional average output current is required.

[0080] In contrast to the disadvantages of the prior art power supply,the power supply of the invention has many advantages. The efficiency ofthe power supply is preferably about 75%. Therefore, for a given powerrequirement of the power supply, the average and peak input currents ofthe power supply of the invention will be significantly lower than thoseof the prior art power supply (e.g., the cat ear power supply). Theselower input currents are especially advantageous when operatingelectronic low voltage transformer type loads. Indeed, even a powersupply with an efficiency of about 50% represents a significantimprovement. Further, the efficiency is reasonably independent of thedifference between the input and output voltage of the power supply.Hence, the power supply of the invention is not limited to operationaround the time of the AC line voltage zero cross as is the prior artcat ear power supply. Indeed, one of the advantages of the power supplyof the invention is the ability to draw input current throughout theduration of the AC line voltage half cycle.

[0081] The power supply of the invention preferably uses a buckconverter configuration to accomplish the stepping down of the voltage.It will be apparent to one having ordinary skill in the art that otherefficient high frequency switching regulators may be employed. Anothersuch configuration is the flyback converter.

[0082] The invention may be embodied in the form of appropriate computersoftware, or in the form of appropriate hardware or a combination ofappropriate hardware and software without departing from the spirit andscope of the present invention. Further details regarding such hardwareand/or software should be apparent to the relevant general public.Accordingly, further descriptions of such hardware and/or softwareherein are not believed to be necessary.

[0083] Although illustrated and described herein with reference tocertain specific embodiments, the present invention is nevertheless notintended to be limited to the details shown. Rather, variousmodifications may be made in the details within the scope and range ofequivalents of the claims and without departing from the invention.

What is claimed is:
 1. An electronic control system operable in a twowire mode and a three wire mode, comprising: a detector having a hotinput terminal and a neutral input terminal and generating at least oneoutput signal, the output signal used to automatically operate theelectronic control system in one of the two wire mode and the three wiremode.
 2. The system of claim 1, wherein the at least one output signalcomprises a hot zero cross detection signal and a neutral zero crossdetection signal, and wherein the detector comprises: a hot zero crossdetector coupled to the hot input terminal to generate the hot zerocross detection signal; and a neutral zero cross detector coupled to theneutral input terminal to generate the neutral zero cross detectionsignal.
 3. The system of claim 1, further comprising a microprocessorcoupled to the detector to monitor the output signal and select one ofthe two wire mode and the three wire mode responsive to the outputsignal.
 4. An electronic control system connectable to a source ofelectric power, operable in a two wire mode and a three wire mode,comprising a hot terminal, a dimmed hot terminal, a neutral terminal anda power supply, the power supply drawing a power supply current from thesource of electric power, wherein said power supply current only flowsbetween the hot terminal and the dimmed hot terminal when saidelectronic control system is operating in said two wire mode, andwherein a portion of said power supply current flows between the hotterminal and neutral terminal when said electronic control system isoperating in said three wire mode.
 5. The power supply of claim 4,wherein the power supply comprises a high frequency switching powersupply.
 6. An electronic control system connectable to a line voltagehaving line voltage zero crossings, comprising a controllably conductivedevice, said electronic control system operable to detect a line voltagezero crossing by causing said controllably conductive device to beconductive for a predetermined period of time prior to said electroniccontrol system monitoring the line voltage for the line voltage zerocrossing.
 7. The system of claim 6, wherein said controllably conductivedevice is controlled to be conductive throughout the monitoring of theline voltage for the line voltage zero crossing.
 8. The system of claim6, wherein the electronic control system is operable in a two wire mode.9. The system of claim 8, wherein the controllably conductive device iscontrolled to be non-conductive prior to said electronic control systemmonitoring the line voltage for the line voltage zero crossing.
 10. Thesystem of claim 6, wherein the predetermined period of time is at leastabout 200 μsec.
 11. The system of claim 10, wherein the monitoring ofthe line voltage for the line voltage zero crossing begins at leastabout 10% of the time between two consecutive line voltage zerocrossings before the line voltage zero crossing.
 12. The system of claim10, wherein the monitoring of the line voltage for the line voltage zerocrossing begins at least about 1 millisecond before the line voltagezero crossing.
 13. The system of claim 12, wherein the controllablyconductive device is controlled to be conductive throughout the timewhen said electronic control system is monitoring the line voltage forthe line voltage zero crossing.
 14. The system of claim 6, wherein theelectronic control system is operable in a three wire mode.
 15. Anelectronic control system comprising at least one controllablyconductive device driven through a high impedance path during fault-freeoperation of said electronic control system and through a low impedancepath after a fault condition has been detected by said electroniccontrol system.
 16. The system of claim 15, further comprising an overvoltage protector that senses an over voltage fault condition present onsaid at least one controllably conductive device and causes said atleast one controllably conductive device to be conductive.
 17. Thesystem of claim 16, further comprising a latching circuit to maintainthe conduction of said at least one controllably conductive device afterthe over voltage fault condition has been cleared.
 18. The system ofclaim 16, further comprising an over current protector that senses anover current fault condition of said at least one controllablyconductive device and causes said at least one controllably conductivedevice to be non-conductive.
 19. The system of claim 18, furthercomprising a lockout circuit which prevents the over voltage protectorfrom controlling the at least one controllably conductive device afteran over current fault condition has been detected.
 20. The system ofclaim 18, further comprising a lockout circuit which prevents the overcurrent protector from controlling the at least one controllablyconductive device after an over voltage fault condition has beendetected.
 21. The system of claim 15, further comprising an over currentprotector that senses an over current fault condition of said at leastone controllably conductive device and causes said at least onecontrollably conductive device to be non-conductive.
 22. The system ofclaim 21, further comprising a latching circuit to maintain thenon-conduction of said at least one controllably conductive device afterthe over current fault condition has been cleared.
 23. The system ofclaim 15, wherein the high impedance path comprises a first path forcontrolling the rate of transition from conduction to non-conduction ofsaid at least one controllably conductive device and a second path forcontrolling the rate of transition from non-conduction to conduction ofsaid at least one controllably conductive device.
 24. The system ofclaim 23, wherein the impedances of said first and second paths areindependent of each other.
 25. The system of claim 15, wherein the lowimpedance path comprises a third path for controlling the rate oftransition from conduction to non-conduction of said at least onecontrollably conductive device and a fourth path for controlling therate of transition from non-conduction to conduction of said at leastone controllably conductive device.
 26. The system of claim 25, whereinthe impedances of said third and fourth paths are independent of eachother.
 27. A device for controlling the amount of power delivered from asource of power to a load comprising: a controllably conductive deviceconnectable between said source and said load; a control circuit forcontrolling said controllably conductive device, responsive to a userinput signal representative of a predetermined amount of power to bedelivered from said source to said load, said control circuit having afirst mode of operation and a second mode of operation; and a detectorcircuit for detecting the presence of an additional input signal andcausing said control circuit to switch from said first mode of operationto said second mode of operation when the presence of said additionalinput signal is detected.
 28. The device of claim 27, wherein thedetector circuit causes a signal derived from said additional inputsignal to be provided to said control circuit when the presence of saidadditional input signal is detected.
 29. A device for controlling theamount of power delivered from a source of power to a load comprising: acontrollably conductive device connectable between said source and saidload, said controllably conductive device having a conductive state anda non-conductive state; a first control circuit for controlling saidcontrollably conductive device in a normal mode of operation responsiveto a user input signal representative of a predetermined amount of powerto be delivered from said source to said load, said first controlcircuit causing said controllably conductive device to transitionbetween said conductive state and said non-conductive state at a firsttransition rate; a second control circuit for controlling saidcontrollably conductive device in a fault mode of operation responsiveto the detection of a fault condition, said second control circuitcausing said controllably conductive device to transition between saidconductive state and said non-conductive state at a second transitionrate which is different from said first transition rate.
 30. The deviceof claim 29, wherein the first transition rate is slower than the secondtransition rate.
 31. The device of claim 29, wherein the firsttransition rate comprises a first turn-on rate and a first turn-off rateand the second transition rate comprises a second turn-on rate and asecond turn-off rate.
 32. The device of claim 31, wherein the firstturn-on rate is different than the second turn-on rate.
 33. The deviceof claim 31, wherein the first turn-off rate is different than thesecond turn-off rate.
 34. An apparatus for controlling the amount ofpower delivered from a source of power to a load comprising: a firstmain terminal and a second main terminal, said first main terminalconnectable to said source of power and said second main terminalconnectable to said load to allow current to flow from said source ofpower to said load; a power supply that draws a power supply currentfrom said source of power through said load; a third terminalconnectable to said source of power, wherein when said third terminal isenergized by said source of power a portion of said power supply currentflows through said third terminal instead of through said load.
 35. Theapparatus of claim 34, wherein said first main terminal is connectableto a hot terminal of said source of power.
 36. The apparatus of claim35, wherein said third terminal is connectable to a neutral connectionof said source of power.
 37. The apparatus of claim 34, furthercomprising a diode that steers said portion of said power supply currentthrough said third terminal instead of through said load.
 38. Anapparatus for controlling the amount of power delivered from a source ofAC power to a load, the AC power having a substantially sinusoidal linevoltage at a predetermined line frequency with zero crossings, theapparatus comprising: a controllably conductive device connectablebetween said source of AC power and said load; and a control circuit forcontrolling the conduction of said controllably conductive device, saidcontrol circuit responsive to an input signal representative of apredetermined amount of power to be delivered from said source of ACpower to said load, said control circuit responsive to said zerocrossings of said substantially sinusoidal line voltage so as tosynchronize the conduction of said controllably conductive device withsaid substantially sinusoidal line voltage; said control circuitenabling a first conduction time of said controllably conductive devicethat is a variable conduction time proportional to said predeterminedamount of power to be delivered from said source of AC power to saidload; said control circuit enabling a second conduction time of saidcontrollably conductive device that is a fixed conduction time in thesame half cycle as said first conduction time, said second conductiontime starting prior to the next zero crossing of said substantiallysinusoidal line voltage and ending at a predetermined time with respectto said next zero crossing; said control circuit causing saidcontrollably conductive device to be non-conductive for a period of timebetween the end of said first conduction time and the beginning of saidsecond conduction time.
 39. The apparatus of claim 38, wherein thesecond conduction time is about 200 μsec.
 40. The apparatus of claim 38,wherein the second conduction time ends at about the time of said nextzero crossing.
 41. A method of reducing flicker in a lamp driven by anelectronic transformer in a system powered by an AC line voltage,comprising the steps of: providing current to said electronictransformer through a series connectable dimming circuit, wherein saidcurrent flows for a user selectable first conduction time in an AC linevoltage half cycle; and providing a non-overlapping second conductiontime in the same half cycle of the AC line voltage just prior to thenext zero crossing of the AC line voltage.
 42. The method of claim 41,wherein said second conduction time is a fixed amount of time.
 43. Themethod of claim 41, wherein said fixed amount of time is about 200microseconds.
 44. The method of claim 41, wherein said second conductiontime ends about 1000 microseconds before said next zero crossing of saidAC line voltage.
 45. A power controlling device for controlling theamount of power delivered from a source of power to a load comprising: afirst and a second main terminal, said first main terminal connectableto said source of power, said second main terminal connectable to saidload to allow current to flow from said source of power to said load;and a power supply that draws a power supply current from said source ofpower and through said load, said power supply having an efficiencygreater than about 50%.
 46. The power controlling device of claim 45,wherein said power supply is a switching type power supply.
 47. Thepower controlling device of claim 46, wherein said power supply is abuck converter type switching supply.
 48. The power controlling deviceof claim 46, wherein said power supply is a flyback type switchingsupply.
 49. The power controlling device of claim 45, further includinga controllably conductive device connected to said first main terminaland said second main terminal, wherein said power supply is operableduring both times of conduction and non-conduction of said controllablyconductive device.
 50. The power controlling device of claim 45, whereinsaid power supply is constrained to run only during selected times ofthe AC line voltage half cycle.
 51. A method for supplying power to thecontrol circuitry of a power control device including at least onecontrollably conductive device connectable to a load in a two wire mode,comprising the steps of: charging a capacitor through said load to apredetermined high voltage when said controllably conductive device isin a non-conductive state; and drawing current from said capacitor usinga converter having a predetermined efficiency to provide a power supplyvoltage for operation of said control circuitry.
 52. The method of claim51, wherein said converter is a switch mode type converter.
 53. Themethod of claim 51, wherein said converter is a flyback type converter.54. The method of claim 51, wherein said converter is at least about 50%efficient.